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mic27600 36v, 7a hyper speed control ? synchronous dc-dc buck regulator superswitcher ii ? hyper speed control, superswitcher ii and any capacitor are trademarks of micrel, inc. mlf and micro leadframe are registered trademarks of amkor technology, inc. micrel inc. ? 2180 fortune drive ? san jose, ca 95131 ? usa ? tel +1 ( 408 ) 944-0800 ? fax + 1 (408) 474-1000 ? http://www.micrel.com general description the micrel mic27600 is a constant-frequency, synchronous buck regulator featuring a unique digitally-modified adaptive on-time control architecture. the mic27600 operates over an input supply range of 4.5v to 36v and provides a regulated output of up to 7a of output current. the output voltage is adjustable down to 0.8v with a guaranteed accuracy of 1%, and the device operates at a switching frequency of 300khz. micrel?s hyper speed control ? architecture allows for ultra- fast transient response while reducing the output capacitance and also makes (high v in )/(low v out ) operation possible. this digitally modified adaptive t on ripple control architecture combines the advantages of fixed-frequency operation and fast transient response in a single device. the mic27600 offers a full suite of protection features to ensure protection of the ic during fault conditions. these include undervoltage lockout to ensure proper operation under power-sag conditions, internal soft-start to reduce inrush current, foldback current limit, ?hiccup? mode short- circuit protection and thermal shutdown. all support documentation can be found on micrel?s web site at: www.micrel.com . features ? hyper speed control ? architecture enables - high delta v operation (v in = 36v and v out = 0.8v) - small output capacitance ? 4.5v to 36v voltage input ? adjustable output from 0.8v to 5.5v (v hsd 28v) ? adjustable output from 0.8v to 3.6v (v hsd 36v) ? 1% fb accuracy ? any capacitor ? stable - zero-esr to high-esr ? 7a output current capabilit y, up to 95% efficiency ? 300khz switching frequency ? internal compensation, 6ms internal soft-start ? foldback current-limit and ?hiccup? mode short-circuit protection ? thermal shutdown ? supports safe startup into a pre-biased load ? ?40 c to +125 c junction temperature range ? 28-pin 5mm 6mm mlf ? package applications ? distributed power systems ? communications/networking infrastructure ? set-top box, gateways and routers ? printers, scanners, graphic cards and video cards ___________________________________________________________________________________________________________ typical application efficiency (v in = 28v) vs. output current 60 65 70 75 80 85 90 95 0123456789 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v july 2011 m9999-070811
micrel, inc. mic27600 july 2011 2 m9999-070811 ordering information part number voltage switching frequency junction temperature range package lead finish mic27600yjl adjustable 300khz ? 40c to + 125 c 28-pin 5mm 6mm mlf ? pb-free pin configuration 28-pin 5mm 6mm mlf ? (yjl) pin description pin number pin name pin function 13, 14, 15, 16, 17, 18, 19 pvin high-side n-internal mosfet dr ain connection (input): the pv in operating voltage range is from 4.5v to 36v. input capacitors between the pvin pi ns and the power ground (pgnd) are required and keep the connection short. 24 en enable (input): a logic level control of the outpu t. the en pin is cmos-compatible. logic high or floating = enable, logic low = shut down. in the off state, the v dd supply current of the device is reduced (typically 0.7ma). do not pull the en pin above the v dd supply. 25 fb feedback (input): input to the tran sconductance amplifier of the cont rol loop. the fb pin is regulated to 0.8v. a resistor divider connecting the feedback to the output is used to adjust the desired output voltage. 26 sgnd signal ground. sgnd must be connected directly to the ground planes. do not route the sgnd pin to the pgnd pad on the top layer, see pcb layout guidelines for details. 27 vdd v dd bias (input): power to the internal referenc e and control sections of the mic27600. the v dd operating voltage range is from 4.5v to 5.5v. a 2.2f ceramic capacitor from the vdd pin to the pgnd pin must be placed next to the ic. v dd must be powered up at the same time or after v in to make the soft-start function correctly. 2, 5, 6, 7, 8, 21 pgnd power ground. pgnd is the ground path for the mic27600 buck converter power stage. the pgnd pin connects to the sources of low-side n-channel internal mosfets, the negative terminals of input capacitors, and the negative terminals of output capacitors. the loop for the power ground should be as small as possible and separate fr om the signal ground (sgnd) loop. 22 cs current sense (input): high current out put driver return. the cs pin connects directly to the switch node. due to the high-speed switching on this pin, t he cs pin should be routed away from sensitive nodes. cs pin also senses the current by moni toring the voltage across the low-side internal mosfet during off-time.
micrel, inc. mic27600 july 2011 3 m9999-070811 pin description (continued) pin number pin name pin function 20 bst boost (output): bootstrappe d voltage to the high-side n-channel internal mosfet driver. a schottky diode is connected between the vdd pin an d the bst pin. a boost capacitor of 0.1 f is connected between the bst pin and the sw pin. 4, 9, 10, 11, 12 sw switch node (output): internal connection for t he high-side mosfet source and low-side mosfet drain. 23 vin power supply voltage (input): requires bypass capacitor to sgnd. 1, 3, 28 nc no connect.
micrel, inc. mic27600 july 2011 4 m9999-070811 absolute maximum ratings (1, 2) pv in to pgnd................................................ ? 0.3v to +38v v in to pgnd .................................................... ? 0.3v to pv in v dd to pgnd ................................................... ? 0.3v to +6v v sw , v cs to pgnd .............................. ? 0.3v to (pv in +0.3v) v bst to v sw ........................................................ ? 0.3v to 6v v bst to pgnd .................................................. ? 0.3v to 44v v en to pgnd ...................................... ? 0.3v to (v dd + 0.3v) v fb to pgnd....................................... ? 0.3v to (v dd + 0.3v) pgnd to sgnd ........................................... ? 0.3v to +0.3v junction temperature .............................................. +150c storage temperature (t s )......................... ? 65 c to +150 c lead temperature (solde ring, 10sec ) ........................ 260c operating ratings (3) supply voltage (pv in , v in )................................. 4.5v to 36v bias voltage (v dd )............................................ 4.5v to 5.5v enable input (v en ) ................................................. 0v to v dd junction temperature (t j ) ........................ ? 40 c to +125 c maximum power dissi pation......................................note 4 package thermal resistance (4) 5mm x 6mm mlf ? ( ja ) ....................................36 c/w electrical characteristics (5) pv in = v in = 12v, v dd = 5v; v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate ? 40c ? t j ? +125c. parameter condition min. typ. max. units power supply input input voltage range (v in , pv in ) 4.5 36 v v dd bias voltage operating bias voltage (v dd ) 4.5 5 5.5 v under-voltage lockout trip level v dd rising 2.4 2.7 3.2 v uvlo hysteresis 50 mv quiescent supply current v fb = 1.5v 1.4 3 ma shutdown supply current v dd = v bst = 5.5v, v in = 36v sw = unconnected, v en = 0v 0.7 2 ma reference 0c ? t j ? 85c (1.0%) 0.792 0.8 0.808 feedback reference voltage ? 40c ? t j ? 125c (1.5%) 0.788 0.8 0.812 v load regulation i out = 0a to 7a 0.2 % line regulation v in = (v out + 3.0v) to 36v 0.1 % fb bias current v fb = 0.8v 5 na dc-dc converter 3.0v ? v hsd ? 28v 0.8 5.5 output voltage range 3.0v ? v hsd ? 36v 0.8 3.6 v enable control en logic level high 4.5v < v dd < 5.5v 1.2 0.85 v en logic level low 4.5v < v dd < 5.5v 0.78 0.4 v en bias current v en = 0v 50 a notes: 1. exceeding the absolute maximum rating may damage the device. 2. devices are esd sensitive. handling pr ecautions recommended. human body model, 1.5k ? in series with 100pf. 3. the device is not guaranteed to function outside operating range. 4. pd (max) = (t j(max) ? t a )/ ja , where ja depends upon the printed circuit layout. see ?applications information.? 5. specification for packaged product only.
micrel, inc. mic27600 july 2011 5 m9999-070811 electrical characteristics (5) (continued) pv in = v in = 12v, v dd = 5v; v bst ? v sw = 5v; t a = 25c, unless noted. bold values indicate ? 40c ? t j ? +125c. parameter condition min. typ. max. units oscillator switching frequency (6) 225 300 375 khz maximum duty cycle (7) v fb = 0v 87 % minimum duty cycle v fb > 0.8v 0 % minimum off-time 360 ns soft-start soft-start time 6 ms short circuit protection current-limit threshold v fb = 0.8v 7.7 15 a short-circuit current v fb = 0v 6 a internal fets top-mosfet r ds (on) i sw = 1a 25 m? bottom-mosfet r ds (on) i sw = 1a 10 m? sw leakage current v in = 36v, v sw = 36v, v en = 0v, v bst = 41v 60 a v in leakage current v in = 36v, v sw = 0v, v en = 0v, v bst = 41v 25 a thermal protection over-temperature shutdown t j rising 155 c over-temperature shutdown hysteresis 10 c notes: 6. measured in test mode. 7. the maximum duty-cycle is limited by the fixed mandatory off-time t off of typically 360ns.
micrel, inc. mic27600 july 2011 6 m9999-070811 typical characteristics v in operating supply current vs. input voltage 0 4 8 12 16 20 5 10152025303540 input voltage (v) supply cur rent (ma) v out = 3.3v i out = 0a v dd = 5v s witc hing v in shutdown current vs. input voltage 0 4 8 12 16 20 5 10152025303540 input voltage (v) shutdown current (a) v dd = 5v v en = 0v v dd operating supply current vs. input voltage 0 2 4 6 8 10 5 10152025303540 input voltage (v) supply current (ma) v out = 3.3v v dd = 5v s witc hing feedback voltage vs. input voltage 0.792 0.796 0.800 0.804 0.808 5 10152025303540 input voltage (v) feedback voltage (v) v out = 3.3v v dd = 5v i out = 0a total regulation vs. input voltage 0.0% 0.2% 0.4% 0.6% 0.8% 1.0% 5 10152025303540 input voltage (v) total regulation (%) v out = 3.3v v dd = 5v i out = 0a to 7a current limit vs. input voltage 0 5 10 15 20 5 10152025303540 input voltage (v) current limit (a) v out = 3.3v v dd = 5v switching frequency vs. input voltage 200 250 300 350 400 5 10152025303540 input voltage (v) switching frequency (khz) v out = 3.3v v dd = 5v i out = 0a v dd operating supply current vs. temperature 0 2 4 6 8 10 -50 -20 10 40 70 100 130 temperature (c) supply current (ma) v in = 28v v out = 3.3v v dd = 5v i out = 0a switching v dd shutdown current vs. temperature 0 0.2 0.4 0.6 0.8 1 -50 -20 10 40 70 100 130 temperature (c) supply current (ma) v in = 28v i out = 0a v dd = 5v v en = 0v
micrel, inc. mic27600 july 2011 7 m9999-070811 typical characteristics (continued) v dd uvlo threshold vs. temperature 2.3 2.4 2.5 2.6 2.7 2.8 -50 -20 10 40 70 100 130 temperature (c) v dd threshold (v) rising falling v in operating supply current vs. temperature 0 4 8 12 16 20 -50 -20 10 40 70 100 130 temperature (c) supply current (ma) v in = 28v v out = 3.3v v dd = 5v i out = 0a s witc hing v in shutdown current vs. temperature 0 4 8 12 16 20 -50 -20 10 40 70 100 130 temperature (c) supply current (a) v in = 28v v dd = 5v i out = 0a current limit vs. temperature 0 5 10 15 20 25 -50 -20 10 40 70 100 130 temperature (c) current limit (a) v in = 28v v out = 3.3v v dd = 5v feedback voltage vs. temperature 0.792 0.796 0.800 0.804 0.808 -50 -20 10 40 70 100 130 temperature (c) feeback voltage (v) v in = 28v v out = 3.3v v dd = 5v i out = 0a load regulation vs. temperature 0.0% 0.2% 0.4% 0.6% 0.8% 1.0% -50 -20 10 40 70 100 130 temperature (c) load regulation (%) v in = 28v v out = 3.3v v dd = 5v i out = 0a to 7a line regulation vs. temperature 0.0% 0.1% 0.2% 0.3% 0.4% 0.5% -50 -20 10 40 70 100 130 temperature (c) line regulation (%) v in = 5.5v to 36v v out = 3.3v v dd = 5v switching frequency vs. temperature 200 250 300 350 400 -50 -20 10 40 70 100 130 temperature (c) switching frequency (khz) v in = 28v v out = 3.3v v dd = 5v i out = 0a en bias current vs. temperature 0 20 40 60 80 100 -50 -20 10 40 70 100 130 temperature (c) en bias current (a) v in = 28v v out = 3.3v v dd = 5v v en = 0v
micrel, inc. mic27600 july 2011 8 m9999-070811 typical characteristics (continued) efficiency vs. output current 50 55 60 65 70 75 80 85 90 95 100 01234567 output current (a) efficiency (%) 12v in 28v in 36v in v out = 3.3v v dd = 5v feedback voltage vs. output current 0.792 0.796 0.800 0.804 0.808 01234567 output current (a) feedback voltage (v) v in = 28v v out = 3.3v v dd = 5v line regulation vs. output current 0.0% 0.1% 0.2% 0.3% 0.4% 0.5% 01234567 output current (a) line regulation (%) v in = 6v to 36v v out = 3.3v v dd = 5v die temperature* (v in = 12v) vs. output current 0 20 40 60 80 01234567 output current (a) die temperature (c) v in = 12v v out = 3.3v v dd = 5v die temperature* (v in = 28v) vs. output current 0 20 40 60 80 01234567 output current (a) die temperature (c) v in = 28v v out = 3.3v v dd = 5v die temperature* (v in = 36v) vs. output current 0 20 40 60 80 01234567 output current (a) die temperature (c) v in = 36v v out = 3.3v v dd = 5v efficiency (v in = 12v) vs. output current 70 75 80 85 90 95 100 0123456789 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v efficiency (v in = 28v) vs. output current 60 65 70 75 80 85 90 95 0123456789 output current (a) efficiency (%) 5.0v 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v efficiency (v in = 36v) vs. output current 50 55 60 65 70 75 80 85 90 95 0123456789 output current (a) efficiency (%) 3.3v 2.5v 1.8v 1.5v 1.2v 1.0v 0.9v 0.8v die temperature* : the temperature measurement was taken at the hottest point on the mic27600 case mounted on a 5 square inch pcb, see thermal measurement section. actual results will depend upon the size of the pcb, ambient temperature and proximity to other he at emitting components.
micrel, inc. mic27600 july 2011 9 m9999-070811 typical characteristics (continued) thermal derating* vs. ambient temperature 0 1 2 3 4 5 6 7 8 85 95 105 115 125 ambient temperature (c) output current (a) v in = 12v v out = 0.8, 1.2, 2.5, 3.3, 5v 2.5v 0.8v 1.2v 3.3v 5v thermal derating* vs. ambient temperature 0 1 2 3 4 5 6 7 8 75 85 95 105 115 125 ambient temperature (c) output current (a) v in = 24v v out = 0.8, 1.2, 2.5, 3.3, 5v 2.5v 0.8v 1.2v 3.3v 5v die temperature* : the temperature measurement was taken at the hottest point on the mic27600 case mounted on a 5 square inch pcb, see thermal measurement section. actual results will depend upon the size of the pcb, ambient temperature and proximity to other he at emitting components.
micrel, inc. mic27600 july 2011 10 m9999-070811 functional characteristics
micrel, inc. mic27600 july 2011 11 m9999-070811 functional characteristics (continued)
micrel, inc. mic27600 july 2011 12 m9999-070811 functional characteristics (continued)
micrel, inc. mic27600 july 2011 13 m9999-070811 functional diagram figure 1. mic27600 block diagram
micrel, inc. mic27600 july 2011 14 m9999-070811 functional description the mic27600 is an adaptive on-time synchronous step-down dc-dc regulator. it is designed to operate over a wide input voltage range from, 4.5v to 36v, and provides a regulated output vo ltage at up to 7a of output current. a digitally modified adaptive on-time control scheme is employed in to obtain a constant switching frequency and to simplify the control compensation. over-current protection is implemented without the use of an external sense resistor. the device includes an internal soft-start function which reduces the power supply input surge current at start-up by controlling the output voltage rise time. theory of operation figure 1 illustrates the block diagram for the control loop of the mic27600. the output voltage is sensed by the mic27600 feedback pin fb via the voltage divider r1 and r2, and compared to a 0.8v reference voltage v ref at the error comparator through a low gain transconductance (g m ) amplifier. if the feedback voltage decreases and the output of the g m amplifier is below 0.8v, then the error comparator will trigger the control logic and generate an on-time period. the on-time period length is predetermined by the ?fixed t on estimation? circuitry: 300khz v v t in out ed) on(estimat = eq. 1 where v out is the output voltage and v in is the power stage input voltage. at the end of the on-time period, the internal high-side driver turns off the high-side mosfet and the low-side driver turns on the low-side mosfet. the off-time period length depends upon the feedback voltage in most cases. when the feedback voltage decreases and the output of the g m amplifier is below 0.8v, the on-time period is triggered and the off-time period ends. if the off-time period determined by the feedback voltage is less than the minimum off-time t off(min) , which is about 360ns, then the mic27600 control logic will apply the t off(min) instead. t off(min) is required to maintain enough energy in the boost capacitor (c bst ) to drive the high- side mosfet. the maximum duty cycle is obtained from the 360ns t off(min) : s s off(min) s max t 360ns 1 t tt d ?= ? = eq. 2 where t s = 1/300khz = 3.33 s. it is not recommended to use mic27600 with a off-time close to t off(min) during steady-state operation. also, as v out increases, the internal ripple injection will increase and reduce the line regulation performance. therefore, the maximum output voltage of the mic27600 should be limited to 5.5v. please refer to ?setting output voltage? subsection in application information for more details. the actual on-time and re sulting switching frequency will vary with the part-to-part variation in the rise and fall times of the internal mosfets, the output load current, and variations in the v dd voltage. also, the minimum t on results in a lower switching frequency in high v in to v out applications, such as 26v to 1.0v. the minimum t on measured on the mic27600 evaluation board is about 184ns. during load transients, the switching frequency is changed due to the varying off-time. to illustrate, the control lo op operation w ill be analyzed in both steady-state and load transient scenarios. for easy analysis, the gain of the g m amplifier is assumed to be 1. with this assumption, the inverting input of the error comparator is the same as the feedback voltage. figure 2 shows the mic27600 control loop timing during steady-state operation. during steady-state, the g m amplifier senses the feedback voltage ripple, which is proportional to the output voltage ripple and the inductor current ripple, to trigger the on-time period. the on- time is predetermined by the t on estimator. the termination of the off-time is controlled by the feedback voltage. at the valley of the feedback voltage ripple, which occurs when v fb falls below v ref , the off period ends and the next on-time period is triggered through the control logic circuitry. figure 2. mic27600 control loop timing
micrel, inc. mic27600 july 2011 15 m9999-070811 figure 3 shows the operation of the mic27600 during a load transient. the output voltage drops due to the sudden load increase, which causes the v fb to be less than v ref . this will cause the error comparator to trigger an on-time period. at the end of the on-time period, a minimum off-time t off(min) is generated to charge c bst since the feedback voltage is still below v ref . then, the next on-time period is triggered due to the low feedback voltage. therefore, the sw itching frequency changes during the load transient, but returns to the nominal fixed frequency once the output has stabilized at the new load current level. with the va rying duty cycle and switching frequency, the output recovery time is fast and the output voltage deviation is small in mic27600 converter. figure 3. mic27600 load transient response unlike true current-mode cont rol, the mic27600 uses the output voltage ripple to trigger an on-time period. the output voltage ripple is proportional to the inductor current ripple if the esr of the output capacitor is large enough. the mic27600 control loop has the advantage of eliminating the need for slope compensation. in order to meet the stability requirements, the mic27600 feedback voltage ripple should be in phase with the inductor current ripple and large enough to be sensed by the g m amplifier and the error comparator. the recommended feedback voltage ripple is 20mv~100mv. if a low-esr output capacitor is selected, then the feedback voltage ripple may be too small to be sensed by the g m amplifier and the error comparator. also, the output voltage ripple and the feedback voltage ripple are not necessarily in phase with the inductor current ripple if the esr of the output capacitor is very low. in these cases, ripple in jection is required to ensure proper operation. please refer to ?ripple injection? subsection in application information for more details about the ripple injection technique. soft-start soft-start reduces the power supply input surge current at startup by cont rolling the output voltage rise time. the input surge appears while the output capacitor is charged up. a slower output ri se time will draw a lower input surge current. the mic27600 implements an internal digital soft-start by making the 0.8v reference voltage v ref ramp from 0 to 100% in about 6ms with 9. 7mv steps. therefore, the output voltage is controlled to increase slowly by a stair- case v fb ramp. once the soft-start cycle ends, the related circuitry is disabled to reduce current consumption. v dd must be powered up at the same time or after v in to make the soft-start function correctly. current limit the mic27600 uses the r ds(on) of the internal low-side power mosfet to sense over -current conditions. this method will avoid adding cost, board space and power losses taken by a discrete current sense resistor. the low-side mosfet is used because it displays much lower parasitic oscillations during switching than the high-side mosfet. in each switching cycle of the mic27600 converter, the inductor current is sensed by monitoring the low-side mosfet in the off period. if the peak inductor current is greater than 15a, then the mic27600 turns off the high-side mosfet and a soft-start sequence is triggered. this mode of operation is called ?hiccup mode? and its purpose is to protect the downstream load in case of a hard short. the current-limit threshold has a foldback characteristic related to the feedback voltage, as shown in figure 4. peak inductor current vs. feedback voltage 0.0 4.0 8.0 12.0 16.0 20.0 0.0 0.2 0.4 0.6 0.8 1.0 feedback voltage (v) peak inductor curent (a) v in = 12v v out = 0v figure 4. mic27600 current limit foldback characteristic
micrel, inc. mic27600 july 2011 16 m9999-070811 internal mosfet gate drive figure 1 (block diagram) shows a bootstrap circuit, consisting of d1 (a schottky diode is recommended) and c bst . this circuit supplies energy to the high-side drive circuit. capacitor c bst is charged, while the low-side mosfet is on, and the voltage on the sw pin is approximately 0v. when the high-side mosfet driver is turned on, energy from c bst is used to turn the mosfet on. as the high-side mosfet turns on, the voltage on the sw pin increases to approximately v in . diode d1 is reverse biased and c bst floats high while continuing to keep the high-side mosfet on. the bias current of the high-side driver is less than 10ma so a 0.1 f to 1 f is sufficient to hold the gate voltage with minimal droop for the power stroke (high-side switching) cycle, i.e. bst = 10ma x 3.33 s/0.1 f = 333mv. when the low-side mosfet is turned back on, c bst is recharged through d1. a small resistor r g , which is in series with c bst , can be used to slow down the turn-on time of the high-side n-channel mosfet. the drive voltage is derived from the v dd supply voltage. the nominal low-side gate drive voltage is v dd and the nominal high-side gate drive voltage is approximately v dd ? v diode , where v diode is the voltage drop across d1. an approximate 30ns del ay between the high-side and low-side driver transitions is used to prevent current from simultaneously flowing unimpeded through both mosfets.
micrel, inc. mic27600 july 2011 17 m9999-070811 application information inductor selection values for inductance, peak, and rms currents are required to select the output inductor. the input and output voltages and the inductance value determine the peak-to-peak inductor ripple current. generally, higher inductance values are used with higher input voltages. larger peak-to-peak ripple currents will increase the power dissipation in the inductor and mosfets. larger output ripple currents will also require more output capacitance to smooth out the larger ripple current. smaller peak-to-peak ripple currents require a larger inductance value and therefore a larger and more expensive inductor. a good compromise between size, loss and cost is to set the inductor ripple current to be equal to 20% of the maximum output current. the inductance value is calculated by equation 3: out(max) sw in(max) out in(max) out i20% f v )v (vv l ? = eq. 3 where: f sw = switching frequency, 300khz 20% = ratio of ac ripple current to dc output current v in(max) = maximum power stage input voltage the peak-to-peak inductor current ripple is: l f v )v (vv i sw in(max) out in(max) out l(pp) ? = eq. 4 the peak inductor current is equal to the average output current plus one half of the peak-to-peak inductor current ripple. i l(pk) =i out(max) + 0.5 i l(pp) eq. 5 the rms inductor current is used to calculate the i 2 r losses in the inductor. 12 i ii 2 l(pp) 2 out(max) l(rms) + = eq. 6 maximizing efficiency requires the proper selection of core material and minimizing the winding resistance. the high frequency operation of the mic27600 requires the use of ferrite materials for a ll but the most cost sensitive applications. lower cost iron powder cores may be used but the increase in core loss will reduce the efficiency of the power supply. this is especially noticeable at low output power. the windi ng resistance decreases efficiency at the higher output current levels. the winding resistance must be minimized although this usually comes at the expense of a larger inductor. the power dissipated in the inductor is equal to the sum of the core and copper losses. at higher output loads, the core losses are usually insignificant and can be ignored. at lower output currents, the core losses can be a significant contributor. core loss information is usually available from the magnetics vendor. copper loss in the inductor is calculated by equation 7: p inductor(cu) = i l(rms) 2 r winding eq. 7 the resistance of the copper wire, r winding , increases with the temperature. the value of the winding resistance used should be at the operating temperature: p winding(ht) = r winding(20c) (1 + 0.0042 (t h ? t 20c )) eq. 8 where: t h = temperature of wire under full load t 20c = ambient temperature r winding(20c) = room temperature winding resistance (usually specified by the manufacturer) output capacitor selection the type of the output capacito r is usually determined by its equivalent series resistance (esr). voltage and rms current capability are two ot her important factors for selecting the output capacitor. recommended capacitor types are ceramic, low-esr aluminum electrolytic, os- con and poscap. the output capacitor?s esr is usually the main cause of the output ripple. the output capacitor esr also affects the control loop from a stability point of view. the maximum value of esr is calculated: l(pp) out(pp) c i v esr out eq. 9 where: ? v out(pp) = peak-to-peak output voltage ripple i l(pp) = peak-to-peak inductor current ripple
micrel, inc. mic27600 july 2011 18 m9999-070811 the total output ripple is a combination of the esr and output capacitance. the total ripple is calculated in equation 10: ( ) 2 c l(pp) 2 sw out l(pp) out(pp) out esr i 8fc i v + ? ? ? ? ? ? ? ? = eq. 10 where: d = duty cycle c out = output capacitance value f sw = switching frequency as described in the ?theory of operation? subsection in functional description , the mic27600 requires at least 20mv peak-to-peak ripple at the fb pin to make the g m amplifier and the error comparator behave properly. also, the output voltage ripple should be in phase with the inductor current. therefor e, the output voltage ripple caused by the output capacitors value should be much smaller than the ripple caused by the output capacitor esr. if low-esr capacitors, such as ceramic capacitors, are selected as the output capacitors, a ripple injection method should be applied to provide the enough feedback voltage ripple. please refer to the ?ripple injection? subsecti on for more details. the voltage rating of the capacitor should be 20% greater for aluminum electrolytic or os-con. the output capacitor rms current is calculated in equation 11: 12 i i l(pp) (rms) c out = eq. 11 the power dissipated in the output capacitor is: out out out c 2 (rms) c) diss(c esr i p = eq. 12 input capacitor selection the input capacitor for the power stage input v in should be selected for ripple current rating and voltage rating. tantalum input capacitors may fail when subjected to high inrush currents, caused by turning the input supply on. a tantalum input capacitor?s voltage rating should be at least two times the maximum input voltage to maximize reliability. aluminum electrolytic, os-con, and multilayer polymer film capacitors can handle the higher inrush currents without voltage de-rating. the input voltage ripple will primarily depend on the input capacitor?s esr. the peak input current is equal to the peak inductor current, so: v in = i l(pk) c esr eq. 13 the input capacitor must be rated for the input current ripple. the rms value of input capacitor current is determined at the maximum output current. assuming the peak-to-peak inductor current ripple is low: d)(1d ii out(max) cin(rms) ? eq. 14 the power dissipated in the input capacitor is: p diss(cin) = i cin(rms) 2 c esr eq. 15 ripple injection the v fb ripple required for proper operation of the mic27600 g m amplifier and error comparator is 20mv to 100mv. however, the output voltage ripple is generally designed as 1% to 2% of the output voltage. for a low output voltage, such as a 1v, the output voltage ripple is only 10mv to 20mv, and the feedback voltage ripple is less than 20mv. if the feedback voltage ripple is so small that the g m amplifier and error comparator can?t sense it, then the mic27600 will lose control and the output voltage is not regulated. in order to have some amount of v fb ripple, a ripple injection method is applied for low output voltage ripple applications. the applications are divided into three situations according to the amount of the feedback voltage ripple: 1) enough ripple at the feedback voltage due to the large esr of the output capacitors. as shown in figure 5a, the converter is stable without any ripple injection. the feedback voltage ripple is: (pp) l c fb(pp) i esr r2r1 r2 v out + = eq. 16 where i l(pp) is the peak-to-peak value of the inductor current ripple. 2) inadequate ripple at the feedback voltage due to the small esr of the output capacitors. the output voltage ripple is fed into the fb pin through a feedforward capacitor c ff in this situation, as shown in figure 5b. the typical c ff value is between 1nf and 100nf.
micrel, inc. mic27600 july 2011 19 m9999-070811 with the feedforward capacitor, the feedback voltage ripple is very close to the output voltage ripple: (pp) l fb(pp) iesr v eq. 17 3) virtually no ripple at the fb pin voltage due to the very low esr of the output capacitors. figure 5a. enough ripple at fb figure 5b. inadequate ripple at fb figure 5c. invisible ripple at fb in this situation, the output voltage ripple is less than 20mv. therefore, additional ripple is injected into the fb pin from the switching node sw via a resistor r inj and a capacitor c inj , as shown in figure 5c. the injected ripple is: = sw divin fb(pp) f 1 d)-(1dkv v eq. 18 r1//r2 r r1//r2 k inj div + = eq. 19 where v in = power stage input voltage d = duty cycle f sw = switching frequency 2 = (r1//r2//r inj ) c ff in equations 18 and 19, it is assumed that the time constant associated with c ff must be much greater than the switching period: 1 t f 1 sw <<= ? eq. 20 if the voltage divider resistors r1 and r2 are in the k ? range, a c ff of 1nf to 100nf can easily satisfy the large time constant requirements. also, a 100nf injection capacitor c inj is used in order to be considered as short for a wide range of the frequencies. the process of sizing the ripple injection resistor and capacitors is: step 1. select c ff to feed all output ripples into the feedback pin and make sure the large time constant assumption is satisfied. typical choice of c ff is 1nf to 100nf if r1 and r2 are in k ? range. step 2. select r inj according to the expected feedback voltage ripple using equation 21: d)(1d f v v k sw in fb(pp) div ? = eq. 21 then the value of r inj is obtained as: 1) k 1 ((r1//r2) r div inj ? = eq. 22 step 3. select c inj as 100nf, which could be considered as short for a wide range of the frequencies.
micrel, inc. mic27600 july 2011 20 m9999-070811 setting output voltage the mic27600 requires two resistors to set the output voltage as shown in figure 6. figure 7. internal ripple injection figure 6. voltage-divider configuration thermal measurements measuring the ic?s case temperature is recommended to ensure it is within its operating limits. although this might seem like a very elementary task, it is easy to get erroneous results. the most common mistake is to use the standard thermal couple that comes with a thermal meter. this thermal couple wire gauge is large, typically 22 gauge, and behaves like a heatsink, resulting in a lower case measurement. the output voltage is determined by equation 23: ) r2 r1 (1vv fb out += eq. 23 where, v fb = 0.8v. a typical value of r1 can be between 3k ? and 10k ? . if r1 is too large, it may allow noise to be introduced into the voltage feedback loop. if r1 is too small, it will decrease the efficiency of the power supply, especially at light loads. once r1 is selected, r2 can be calculated using: fb out fb vv r1v r2 ? = eq. 24 two methods of temperature measurement are using a smaller thermal couple wire or an infrared thermometer. if a thermal couple wire is used, it must be constructed of 36 gauge wire or higher then (smaller wire size) to minimize the wire heat-sinking effect. in addition, the thermal couple tip must be covered in either thermal grease or thermal glue to make sure that the thermal couple junction is making good contact with the case of the ic. omega brand thermal couple (5sc-tt-k-36-36) is adequate for most applications. wherever possible, an infrared thermometer is recommended. the measurement spot size of most infrared thermometers is too large for an accurate reading on a small form factor ics. however, a ir thermometer from optris has a 1mm spot size, which makes it a good choice for measuring the hottest point on the case. an optional stand makes it easy to hold the beam on the ic for long periods of time. in addition to the external ripple injection added at the fb pin, internal ripple injection is added at the inverting input of the comparator in side the mic27600, as shown in figure 7. the inverting input voltage v inj is clamped to 1.2v. as v out increases, the swing of v inj will be clamped. the clamped v inj reduces the line regulation because it is reflected back as a dc error on the fb terminal. to avoid this line regulation problem, the maximum output voltage of the mic27600 should be limited to 5.5v for up to 28v v in and 3.6v for v in higher than 28v.
micrel, inc. mic27600 july 2011 21 m9999-070811 pcb layout guidelines warning!!! to minimize emi and output noise, follow these layout recommendations. pcb layout is critical to achieve reliable, stable and efficient performance. a ground plane is required to control emi and minimize the inductance in power, signal and return paths. the following guidelines should be followed to insure proper operation of the mic27600 converter. ic ? the 2.2f ceramic capacitor, which is connected to the vdd pin, must be located right at the ic. the vdd pin is very noise sens itive and placement of the capacitor is very critical. use wide traces to connect to the vdd and pgnd pins. ? the signal ground pin (sgnd) must be connected directly to the ground planes. do not route the sgnd pin to the pgnd pad on the top layer. ? place the ic close to the point of load (pol). ? use fat traces to route the input and output power lines. ? signal and power grounds should be kept separate and connected at only one location. input capacitor ? place the input capacitor next. ? place the input capacitors on the same side of the board and as close to the ic as possible. ? keep both the pvin pin and pgnd connections short. ? place several vias to the ground plane close to the input capacitor ground terminal. ? use either x7r or x5r dielectric input capacitors. do not use y5v or z5u type capacitors. ? do not replace the ceramic input capacitor with any other type of capacitor. any type of capacitor can be placed in parallel with the input capacitor. ? if a tantalum input capacitor is placed in parallel with the input capacitor, it must be recommended for switching regulator applications and the operating voltage must be derated by 50%. ? in ?hot-plug? applications, a tantalum or electrolytic bypass capacitor must be used to limit the over- voltage spike seen on the input supply with power is suddenly applied. inductor ? keep the inductor connection to the switch node (sw) short. ? do not route any digital lines underneath or close to the inductor. ? keep the switch node (sw) away from the feedback (fb) pin. ? the cs pin should be connected directly to the sw pin to accurate sense the voltage across the low- side mosfet. ? to minimize noise, place a ground plane underneath the inductor. ? the inductor can be placed on the opposite side of the pcb with respect to the ic. it does not matter whether the ic or inductor is on the top or bottom as long as there is enough air flow to keep the power components within their temperature limits. the input and output capacitors must be placed on the same side of the board as the ic. output capacitor ? use a wide trace to connect the output capacitor ground terminal to the input capacitor ground terminal. ? phase margin will change as the output capacitor value and esr changes. contact the factory if the output capacitor is different from what is shown in the bom. ? the feedback trace should be separate from the power trace and connected as close as possible to the output capacitor. sensing a long high current load trace can degrade the dc load regulation. rc snubber ? place the rc snubber on either side of the board and as close to the sw pin as possible.
micrel, inc. mic27600 july 2011 22 m9999-070811 evaluation board schematic figure 8. schematic of mic27600 evaluation board (j13, r13, r15 are for testing purposes)
micrel, inc. mic27600 july 2011 23 m9999-070811 bill of materials item part number manufacturer description qty. c1 b41125a6107m epcos (1) 100f aluminum capacitor, smd, 50v 1 12105c475kaz2a avx (2) c2, c3 grm32er71h475ka88l murata (3) 4.7f ceramic capacitor, x7r, size 1210, 50v 2 12106d107mat2a avx c13 grm32er60j107me20l murata 100f ceramic capacitor, x5r, size 1210, 6.3v 1 06035c104kat2a avx grm188r71h104ka93d murata c6, c7, c10 c1608x7r1h104k tdk (4) 0.1f ceramic capacitor, x7r, size 0603, 50v 3 0805zc225mat2a avx grm21br71a225ka01l murata c8, c9 c2012x7r1a225k tdk 2.2f ceramic capacitor, x7r, size 0805, 10v 2 06035c102kat2a avx grm188r71h102ka01d murata c11 c1608x7r1h102k tdk 1nf ceramic capacitor, x7r, size 0603, 50v 1 06035c223kaz2a avx grm188r71h223k murata c12 c1608x7r1h223k tdk 22nf ceramic capacitor, x7r, size 0603, 50v 1 c4, c5 open c14, c15 open sd101aws-7 diodes inc (6) d1 sd101aws-v vishay (7) small signal schottky diode 1 d2 cmdz5l6 central semi (8) 5.6v zener diode 1 l1 hcf1305-4r0-r cooper bussmann (9) 4.0h inductor, 15a saturation current 1 q1 fcx619 zetex 50v npn transistor 1 r1 crcw06034r75fkea vishay dale 4.75 ? resistor, size 0603, 1% 1 r2, r16 crcw08051r21fkea vishay dale 1.21? resistor, size 0805, 1% 2 r3, r4 crcw060310k0fkea vishay dale 10k ? resistor, size 0603, 1% 2 r5 crcw060380k6fkea vishay dale 80.6k ? resistor, size 0603, 1% 1 r6 crcw060340k2fkea vishay dale 40.2k ? resistor, size 0603, 1% 1 r7 crcw060320k0fkea vishay dale 20k ? resistor, size 0603, 1% 1 notes: 1. epcos: www.epcos.com . 2. avx: www.avx.com . 3. murata: www.murata.com . 4. tdk: www.tdk.com . 5. sanyo: www.sanyo.com . 6. diode inc.: www.diodes.com . 7. vishay: www.vishay.com . 8. central semi: www.centralsemi.com . 9. cooper bussmann: www.cooperbussmann.com .
micrel, inc. mic27600 july 2011 24 m9999-070811 bill of materials (continued) item part number manufacturer description qty. r8 crcw060311k5fkea vishay dale 11.5k ? resistor, size 0603, 1% 1 r9 crcw06038k06fkea vishay dale 8.06k ? resistor, size 0603, 1% 1 r10 crcw06034k75fkea vishay dale 4.75k ? resistor, size 0603, 1% 1 r11 crcw06033k24fkea vishay dale 3.24k ? resistor, size 0603, 1% 1 r12 crcw06031k91fkea vishay dale 1.91k ? resistor, size 0603, 1% 1 r13 crcw06030000fkea vishay dale 0 ? resistor, size 0603, 5% 1 r14 crcw06035k23fkea vishay dale 5.23k ? resistor, size 0603, 1% 1 r15 crcw060349r9fkea vishay dale 49.9 ? resistor, size 0603, 1% 1 u1 mic27600yjl micrel. inc. (10) 26v/7a synchronous buck dc-dc regulator 1 note: 10. micrel, inc.: www.micrel.com .
micrel, inc. mic27600 july 2011 25 m9999-070811 pcb layout figure 9. mic27600 evaluation board top layer figure 10. mic27600 evaluation board mid-layer 1 (ground plane)
micrel, inc. mic27600 july 2011 26 m9999-070811 pcb layout (continued) figure 11. mic27600 evaluation board mid-layer 2 figure 12. mic27600 evaluation board bottom layer
micrel, inc. mic27600 july 2011 27 m9999-070811 recommended land pattern
micrel, inc. mic27600 july 2011 28 m9999-070811 package information 28-lead 5mm x 6mm mlf ? (yjl) micrel, inc. 2180 fortune drive san jose, ca 95131 usa tel +1 (408) 944-0800 fax +1 (408) 474-1000 web http://www.micrel.com micrel makes no representations or warranties with respect to t he accuracy or completeness of the information furnished in this data sheet. this information is not intended as a warranty and micrel does not assume responsibility for it s use. micrel reserves the right to change circuitry, specifications and descriptions at any time without notice. no license, whether expre ss, implied, arising by estoppel or other wise, to any intellectual property rights is granted by this document. except as provided in micrel?s terms and conditions of sale for such products, mi crel assumes no liability whatsoever, and micrel disclaims any express or implied warranty relating to the sale and/or use of micrel products including l iability or warranties relating to fitness for a particular purpose, merchantability, or infringement of an y patent, copyright or other intellectual p roperty right micrel products are not designed or authori zed for use as components in life support app liances, devices or systems where malfu nction of a product reasonably be expected to result in pers onal injury. life support devices or system s are devices or systems that (a) are in tended for surgical impla into the body or (b) support or sustain life, and whose failure to perform can be reasonably expected to result in a significan t injury to the user. a purchaser?s use or sale of micrel produc ts for use in life support app liances, devices or systems is a purchaser?s own risk and purchaser agrees to fully indemnify micrel for any damages resulting from such use or sale. can nt ? 2010 micrel, incorporated.

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